Autozeroing current feedback instrumentation amplifier

ABSTRACT

An embodiment is directed to an instrumentation amplifier. The instrumentation amplifier includes an output stage for generating an output voltage, a low-frequency path coupled with the output stage, and a high-frequency path coupled with the output stage. The high-frequency path dominates the low-frequency path at frequencies above a particular frequency, and the low-frequency path dominates the high-frequency path at frequencies below the particular frequency. The low-frequency path includes an input stage for sensing a differential input and generating an intermediate current based thereon, a feedback stage coupled with the input and output stages, the feedback stage for generating a feedback current based on the output voltage, and an auto-zeroing circuit coupled with the input, feedback, and output stages, the auto-zeroing circuit for generating a nulling current. The nulling current compensates for errors in the intermediate and feedback currents resulting from input offsets in the input and feedback stages.

REFERENCE TO CO-PENDING APPLICATIONS FOR PATENT

The present Application for Patent is related to co-pending U.S. patentapplication Ser. No. 11/804,490, now U.S. Pat. No. 7,573,327, entitled“AUTOZEROING CURRENT FEEDBACK INSTRUMENTATION AMPLIFIER”, by MichielPertijs et al., filed May 17, 2007, assigned to the assignee hereof, andexpressly incorporated by reference herein.

BACKGROUND

1. Field

Embodiments generally relate to the field of current feedbackinstrumentation amplifiers.

2. Background

Instrumentation amplifiers are commonly used to amplify smalldifferential input voltages while rejecting common-mode input voltages.A desired feature of such amplifiers is a low input-referred offsetvoltage combined with a low input current. The latter can be achieved byusing a MOS input stage, but such an input stage typically results in ahigh offset voltage.

Another desired feature of instrumentation amplifiers is that theirinput range includes the negative supply rail, so that they can beconnected to a grounded signal source in a single-supply system. This isnot possible with a conventional 3-opamp instrumentation amplifiertopology.

This limitation to some extent has been overcome by using acurrent-feedback topology with PMOS input transistors. The PMOStransistors transfer the differential input voltage to a resistorconnected between their sources, resulting in a current proportional tothe differential input voltage. The PMOS transistors at the same timeprovide the required common-mode level-shift to be able to make thisvoltage-to-current conversion with an input voltage at ground level. Inthe rest of the amplifier, the generated current is converted back intoan output voltage using a second resistor.

FIG. 1 shows a block diagram of one conventional current-feedbackinstrumentation amplifier 100. The differential input voltage V_(in) isconverted to a current by transconductance amplifier g₂. As describedabove, amplifier g₂ has PMOS input transistors which enable it to senseinput signals at the negative supply rail. The difference between theoutput voltage V_(out) and a reference voltage V_(ref) is scaled down bya resistive divider, consisting of R₁ and R₂, to provide a feedbackvoltage V_(fb). This is applied to a second transconductance amplifierg₃. The feedback loop, closed by the output stage g₁, ensures that theoutput current of g₃ equals that of g₂. It is appreciated that theoutput stage, here shown as a single Miller-compensated transconductancestage, can in practice consist of multiple stages. If the twotransconductances g₂ and g₃ are equal, V_(fb) equals V_(in), andtherefore the output voltage equals:V _(out) =V _(ref)+(R ₁ +R ₂)/R ₂ ·V _(in)  (1)

In the more general case that the two transconductances are not equal,the output voltage equals:V _(out) =V _(ref) +g ₂ /g ₃·(R ₁ +R ₂)/R ₂ ·V _(in)  (2)

In addition to its ability to sense input voltages at the negativesupply rail, amplifier 100 has the attractive feature that its outputcan swing rail-to-rail, which is important in low-voltage applications.

However, circuit 100 is disadvantageous in that the offsets oftransconductance amplifiers g₂ and g₃ add directly to the input voltage,and therefore need to be compensated for. FIG. 2 illustrates oneconventional amplifier 200 that employs chopper switches 210 and 220added at the input of g₂ and g₃ to periodically reverse the polarity ofthe input and feedback signal. An additional chopper switch 230 at theinput of the output stage restores the original polarity. Thisconfiguration effectively modulates the offset of the transconductanceamplifiers to the chopper frequency, where it can, in principle, befiltered out.

An important disadvantage of using chopping to eliminate the offset incurrent-feedback instrumentation amplifiers is that the modulated offsetresults in spurious AC signals at the output of the amplifier 200. Forexample, the output of amplifier 200 may actually appear as a sawtoothsignal. Since the output of an instrumentation amplifier is typicallysampled by an analog-to-digital converter, such spurious signal mayresult in measurement errors unless they are filtered out. Conventionalimplementations have attempted to reduce and filter these spurioussignals by using a continuous (non-chopped) feedforward path and variousextra offset-compensation loops. This, however, leads to a very largeand complex system.

Another important disadvantage of using chopping is that the inputsource is exposed to a switched capacitive load consisting of the inputcapacitance C_(in2) of transconductance amplifier g₂. Due to theperiodic polarity reversal, this input capacitance has to be alternatelycharged to +V_(in) and −V_(in). The associated current results in aninput offset current. Effectively, this reduces the input impedance ofthe instrumentation amplifier (e.g., amplifier 200) to:R _(in)=1/(2·f _(chop) ·C _(in2)).  (3)For typical values of f_(chop)=10 kHz and C_(in2)=1 pF, the inputimpedance is 50 MΩ. In contrast, non-chopped instrumentations amplifierswith MOS inputs typically achieve input impedances on the order of 10GΩ. This reduced impedance due to chopping can cause significant gainerrors when reading out a high-impedance signal source. A similarproblem occurs at the input of transconductance amplifier g₃, whoseinput capacitance C_(in3) presents a switched load to the feedbacknetwork.

Thus, conventional current feedback instrumentation amplifiers do notprovide a simple way to reduce input offsets while at the same timemaintaining high input impedance and avoiding spurious signals at theoutput.

SUMMARY

This summary is provided to introduce a selection of concepts in asimplified form that are further described below in the DetailedDescription. This summary is not intended to identify key features oressential features of the claimed subject matter, nor is it intended tobe used to limit the scope of the claimed subject matter.

An embodiment is directed to an instrumentation amplifier. Theinstrumentation amplifier includes an output stage for generating anoutput voltage, a low-frequency path coupled with the output stage, anda high-frequency path coupled with the output stage. The high-frequencypath dominates the low-frequency path at frequencies above a particularfrequency, and the low-frequency path dominates the high-frequency pathat frequencies below the particular frequency. The low-frequency pathincludes an input stage for sensing a differential input and generatingan intermediate current based thereon, a feedback stage coupled with theinput and output stages, the feedback stage for generating a feedbackcurrent based on the output voltage, and an auto-zeroing circuit coupledwith the input, feedback, and output stages, the auto-zeroing circuitfor generating a nulling current. The nulling current compensates forerrors in the intermediate and feedback currents resulting from inputoffsets in the input and feedback stages.

Thus, embodiments provide technology allowing for instrumentationamplifiers with very low input-referred offset, low input current, andlow level spurious switching signals at the output. Additionally,spurious signals may be further reduced by adding a high-frequencyfeedforward path.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated in and form a part ofthis specification, illustrate embodiments of the invention and,together with the description, serve to explain the principles ofembodiments of the invention:

FIG. 1 shows a block diagram of one conventional current-feedbackinstrumentation amplifier.

FIG. 2 illustrates one conventional amplifier that employs chopperswitches added at the input and output of g₂ and g₃ from FIG. 1 tomodulate the offset of g₂ and g₃ away from DC.

FIG. 3 illustrates a block diagram of a current feedback amplifier, inaccordance with various embodiments of the present invention.

FIG. 4 illustrates a schematic of a current feedback amplifier, inaccordance with various embodiments of the present invention.

FIG. 5 illustrates a block diagram of a current feedback amplifier,including a high-frequency feedforward path, in accordance with variousembodiments of the present invention.

FIG. 6 illustrates a schematic of a current feedback instrumentationamplifier, including a high-frequency feedforward path, in accordancewith various embodiments of the present invention.

FIG. 7 illustrates a schematic of a current feedback instrumentationamplifier, including parallel input stages, in accordance with variousembodiments of the present invention.

FIG. 8 illustrates a schematic of a current feedback instrumentationamplifier, including parallel input stages, in accordance with variousembodiments of the present invention.

FIG. 9 illustrates a block diagram of a current feedback instrumentationamplifier, including parallel input stages and a high-frequencyfeedforward path, in accordance with various embodiments of the presentinvention.

FIG. 10 illustrates a schematic of a current feedback instrumentationamplifier, including parallel input stages and a high-frequencyfeedforward path, in accordance with various embodiments of the presentinvention.

FIG. 11 illustrates a schematic of a current feedback instrumentationamplifier, including a current buffer stage, in accordance with variousembodiments of the present invention.

FIG. 12 illustrates an input stage of an instrumentation amplifier thatincludes pre-charging circuitry, in accordance with various embodimentsof the present invention.

FIG. 13 illustrates a flowchart of a process for reducing effects ofoffsets in a current feedback instrumentation amplifier, in accordancewith various embodiments of the present invention.

FIG. 14 illustrates a flowchart of a process for generating a nullingcurrent, in accordance with various embodiments of the presentinvention.

FIG. 15 illustrates a flowchart of a process for switching aninstrumentation amplifier from an amplification configuration to anauto-zero configuration, in accordance with various embodiments of thepresent invention.

FIGS. 16A-16B illustrate a flowchart for a process for reducing theeffects of offsets in an instrumentation amplifier, in accordance withvarious embodiments of the present invention.

FIG. 17 illustrates a flowchart for a process of calibrating a nullingcurrent, in accordance with various embodiments of the presentinvention.

DETAILED DESCRIPTION

Reference will now be made in detail to the preferred embodiments of theinvention, examples of which are illustrated in the accompanyingdrawings. While the invention will be described in conjunction with thepreferred embodiments, it will be understood that they are not intendedto limit the invention to these embodiments. On the contrary, theinvention is intended to cover alternatives, modifications andequivalents, which may be included within the spirit and scope of theinvention as defined by the claims. Furthermore, in the detaileddescription of the present invention, numerous specific details are setforth in order to provide a thorough understanding of the presentinvention. However, it will be obvious to one of ordinary skill in theart that the present invention may be practiced without these specificdetails. In other instances, well known methods, procedures, components,and circuits have not been described in detail as not to unnecessarilyobscure aspects of the present invention.

Overview

Generally speaking, embodiments provide technology for reducing inputoffsets in current feedback instrumentation amplifiers. The technologyinvolves using auto-zeroing circuitry to null an offset of an inputstage. In one embodiment, this is achieved by periodically switching-inthe auto-zeroing circuitry. As a result, embodiments are able to achievevery low input-referred offset, low input current, and low-levelspurious switching signals at the output. Additionally, spurious signalsmay be further reduced by adding a high-frequency feedforward path.

Exemplary Circuits, in Accordance with Various Embodiments

FIG. 3 illustrates a block diagram of a current feedback amplifier 300,in accordance with various embodiments of the present invention.Amplifier 300 includes an input stage 320, an output stage 310, afeedback stage 330, and a feedback network 350. The feedback network isoperable to generate a feedback voltage V_(fb) from the output voltageV_(out) and the reference voltage V_(ref) and thus defines the gain ofthe amplifier. The amplifier 300 also advantageously includes anauto-zero circuit 340 coupled to the input stage 320, the output stage310, and the feedback stage 330. The auto-zero circuit 340 is operableto switch the amplifier 300 between an amplification configurationcorresponding to an amplification phase and an auto-zeroingconfiguration corresponding to an auto-zeroing phase. During theamplification phase, the amplifier 300 is operable to perform normalamplification operations. During the auto-zeroing phase, the auto-zerocircuit 340 is operable to null offset currents generated by the inputstage 320 and the feedback stage 330.

In one embodiment, the auto-zero circuit 340 nulls the offset currentsby shorting inputs of the input stage 320 and the feedback stage 330 torespective common mode voltages. Subsequently, the auto-zero circuit 340may then measure corresponding offset currents generated by the inputstage 320 and the feedback stage 330 and generate a nulling currentbased thereon. The nulling current serves to compensate for the offsetcurrents generated by the input stage 320 and the feedback stage 330.

When the auto-zero circuit 340 switches the amplifier 300 back to theamplification configuration, the auto-zero circuit 340 continues togenerate the nulling current, thereby reducing or even eliminatingoffsets in the amplifier 300. In one embodiment, the auto-zero circuit340 periodically switches between amplification and auto-zeroingconfigurations so as to periodically recalibrate the nulling current.

It should be appreciated that amplifier 300 may be achieved in a numberof ways. For example, FIG. 4 illustrates a schematic of a currentfeedback amplifier 400, in accordance with various embodiments of thepresent invention. In amplifier 400, tranconductance amplifier 411serves as an output stage, such as output stage 310 in amplifier 300,tranconductance amplifier 412 serves as an input stage, such as inputstage 320 of amplifier 300, tranconductance amplifier 413 serves as afeedback stage, such as feedback stage 330 of amplifier 300, andresistors 461 and 462 serve as feedback network, such as feedbacknetwork 350 of amplifier 300. Although embodiments may be describedherein with reference to single tranconductance amplifiers, it should beappreciated that embodiments are not limited as such. For example, acascade of multiple stages with appropriate frequency compensation maybe used in place of the single tranconductance amplifier 411 in order toobtain a higher gain. Capacitors 451 and 452 serve as frequencycompensators for the tranconductance amplifier 411, thus forming aMiller-compensated output stage. Resistors 461 and 462, together withthe reference voltage V_(ref), generate the feedback voltage V_(fb)based on the output voltage V_(out). V_(fb) is fed back as an input tothe feedback tranconductance amplifier 413.

Switches 431-436 and 441-446, tranconductance amplifiers 414 and 415,and capacitors 453 and 454 function together as an auto-zero circuit,such as auto-zero circuit 340 of amplifier 300. It should be appreciatedthat switches 431-436 and 441-446 may be any of a number of devicescapable of performing a switching function. In one embodiment, theswitches 431-436 and 441-446 serve to switch the amplifier 400 betweenamplification and auto-zeroing configurations. For example, anamplification configuration may correspond to switches 441-446 beingclosed and switches 431-436 being open. Conversely, an auto-zeroingconfiguration may correspond to switches 431-436 being closed andswitches 441-446 being open.

During the auto-zeroing phase, the inputs of the tranconductanceamplifiers 412 and 413 are respectively shorted to the input common modevoltage V_(cmin) and the feedback common mode voltage V_(cmfb). Anyinput offsets of amplifiers 412 and 413 cause an offset current thatflows into the integrator formed by tranconductance amplifier 414 andcapacitors 453 and 454. The output of this integrator then drives thetranconductance amplifier 415 to generate a nulling current, whicheffectively nulls the offset current.

At the end of the auto-zeroing phase, switches 431-436 open. As aresult, the voltage at the output of the integrator around amplifier 414is held so that amplifier 415 continues nulling the offset current atthe outputs of amplifiers 412 and 413.

Subsequently, in the amplification phase, switches 441-446 are closed.V_(in) and V_(fb) are applied to amplifiers 412 and 413, respectively,and the summed output current of amplifiers 412, 413, and 415 is coupledwith the output stage (i.e., amplifier 411, etc.). The amplifier 400then operates similar to a traditional current feedback instrumentationamplifier, except that the nulling current injected by amplifier 415ensures that the input-referred offset voltages of amplifiers 412-413 donot contribute to the output voltage. Thereafter, in a subsequentauto-zeroing phase, the Miller-compensated output stage formed byamplifier 411 and capacitors 451-452 holds the output voltage whileamplifiers 412-413 are auto-zeroed again.

In some instances, the gating of the input signal may result indetection of components of the input signal (including noise) atharmonics of the clock frequency. Such components may mix with the clocksignal and be modulated down to baseband. Consequently, this may resultin errors and increased noise at the output of amplifiers 300 and 400.

In one embodiment, this mixing may be prevented by using ahigh-frequency feedforward path. FIG. 5 illustrates a block diagram of acurrent feedback amplifier 500, including a high-frequency feedforwardpath, in accordance with various embodiments of the present invention.Amplifier 500 includes input stages 520 and 570, an output stage 510,feedback stages 530 and 580, and a feedback network 550. The amplifier500 also advantageously includes an auto-zero circuit 540 coupled to theinput stage 520, the output stage 510, and the feedback stage 530. Theauto-zero circuit 540 is operable to switch the amplifier 500 between anamplification configuration corresponding to an amplification phase andan auto-zeroing configuration corresponding to an auto-zeroing phase.During the amplification phase, the amplifier 500 is operable to performnormal amplification operations. During the auto-zeroing phase, theauto-zero circuit 540 is operable to null offset currents generated bythe input stage 520 and the feedback stage 530.

In one embodiment, the auto-zero circuit 540 nulls the offset currentsby shorting inputs of the input stage 520 and the feedback stage 530 torespective common mode voltages. Subsequently, the auto-zero circuit 540may then measure corresponding offset currents generated by the inputstage 520 and the feedback stage 530 and generate a nulling currentbased thereon. The nulling current serves to compensate for the offsetcurrents generated by the input stage 520 and the feedback stage 530.

When the auto-zero circuit 540 switches the amplifier 500 back to theamplification configuration, the auto-zero circuit 540 continues togenerate the nulling current, thereby reducing or even eliminatingoffsets in the amplifier 500. In one embodiment, the auto-zero circuit540 periodically switches between amplification and auto-zeroingconfigurations so as to periodically recalibrate the nulling current.

For low frequencies (e.g., below the clock frequency), the auto-zeropath comprising input stage 520, feedback stage 530, and auto-zerocircuit 540 is dominant, and the amplifier 500 then operates similar toa traditional current feedback instrumentation amplifier, except thatthe nulling current injected by the auto-zero circuit 540 ensures thatthe input-referred offsets of input stage 520 and feedback stage 530 donot contribute to the output voltage. Thereafter, in a subsequentauto-zeroing phase, the output stage 510 may hold the output voltagewhile the input stage 520 and the feedback stage 530 are auto-zeroedagain.

At high frequencies, the feedforward path comprising input stage 570 andfeedback stage 580 is dominant. Above a threshold frequency, thefeedforward path ensures that the feedback signal V_(fb) can track theinput signal V_(in). As a result, even if mixing occurs due to thegating at the inputs of input stage 520 and feedback stage 530, theresulting mixing products cancel.

An additional advantage of the feedforward path is that it attenuatesswitching transients produced by the auto-zeroed input stage. The lowerthe threshold frequency, the higher the relative gain of the feedforwardpath at the clock frequency and its harmonics, and therefore the betterthe attenuation of such switching transients.

It is appreciated that amplifier 500 may be achieved in a number ofways. For example, FIG. 6 illustrates a schematic of a current feedbackinstrumentation amplifier 600, including a high-frequency feedforwardpath, in accordance with various embodiments of the present invention.In amplifier 600, tranconductance amplifiers 611 and 616 together serveas an output stage, such as output stage 510 of amplifier 500,tranconductance amplifiers 612 and 617 serve as input stages, such asinput stages 520 and 570 of amplifier 500, tranconductance amplifiers613 and 618 serve as feedback stages, such as feedback stages 530 and580 of amplifier 500, and resistors 661 and 662 together serve asfeedback network, such as feedback network 550 of amplifier 500.Capacitors 651 and 652 serve as frequency compensators for thetranconductance amplifier 611, thus forming a nested-Miller-compensatedoutput driver stage. Additionally, tranconductance amplifier 616, alongwith capacitors 656-657, serves as a Miller-compensated intermediatestage to amplifier 600. Resistors 661 and 662 together with thereference voltage V_(ref) generate the feedback voltage V_(fb) based onthe output voltage V_(out). V_(fb) is fed back as an input to thefeedback tranconductance amplifiers 613 and 618.

Switches 631-636 and 641-646, tranconductance amplifiers 614 and 615,and capacitors 653 and 654 function together as an auto-zero circuit,such as auto-zero circuit 340 of amplifier 300. It should be appreciatedthat switches 631-636 and 641-646 may be any of a number of devicescapable of performing a switching function. In one embodiment, theswitches 631-636 and 641-646 serve to switch the amplifier 600 betweenamplification and auto-zeroing configurations. For example, anamplification configuration may correspond to switches 641-646 beingclosed and switches 631-636 being open. Conversely, an auto-zeroingconfiguration may correspond to switches 631-636 being closed andswitches 641-646 being open.

During the auto-zeroing phase, the inputs of the tranconductanceamplifiers 612 and 613 are respectively shorted to the input common modevoltage V_(cmin) and the feedback common mode voltage V_(cmfb). Anyinput offsets of amplifiers 612 and 613 causes an offset current thatflows into the integrator formed by tranconductance amplifier 614 andcapacitors 653 and 654. The output of this integrator then drives thetranconductance amplifier 615 to generate a nulling current, whicheffectively nulls the offset current.

At the end of the auto-zeroing phase, switches 631-636 open. As aresult, the voltage at the output of the integrator around amplifier 614is held so that amplifier 615 continues nulling the offset current atthe outputs of amplifiers 612 and 613. Subsequently, in theamplification phase, switches 641-646 are closed. V_(in) and V_(fb) areapplied to amplifiers 612 and 613, respectively, and the summed outputcurrent of amplifiers 612, 613, and 615 is coupled with the intermediatestage (i.e., amplifier 616).

For low frequencies (e.g., below the clock frequency), the auto-zeropath comprising amplifiers 611-616 is dominant, and the amplifier 600then operates similar to a traditional current feedback instrumentationamplifier, except that the nulling current injected by amplifier 615ensures that the input-referred offset voltages of amplifiers 612-613 donot contribute to the output voltage. Thereafter, in a subsequentauto-zeroing phase, the nested-Miller-compensated output stage formed byamplifier 611 and capacitors 651-652 and amplifier 616 and capacitors656-657 hold the output voltage while amplifiers 612-613 are auto-zeroedagain.

At high frequencies, the feedforward path comprising amplifiers 617-618is dominant. The feedforward path, together with the output amplifier611, forms a regular Miller-compensated two-stage amplifier withapproximately 20 dB/dec roll-off. This type of frequency compensation isknown as “multi-path nested-Miller compensation” and has been used inconventional op-amps, but without application to auto-zeroedinstrumentation amplifiers.

In one embodiment, the frequency at which the feedforward path starts todominate is:ω_(pz) =g ₆₁₈ /C ₆₅₁,  (1)(assuming C₆₅₁=C₆₅₂ and g₆₁₇=g₆₁₈). In a preferred embodiment, thisfrequency is chosen to be below the clock frequency. Above ω_(pz), thefeedforward path ensures that the feedback signal V_(fb) can track theinput signal V_(in). As a result, even if mixing occurs due to thegating at the inputs of amplifiers 612-613, the resulting mixingproducts cancel.

The above-referenced mixing problems may alternatively be solved byusing a dual-input-stage “ping-pong” architecture. FIG. 7 illustrates aschematic of a current feedback instrumentation amplifier 700, includingparallel input stages, in accordance with various embodiments of thepresent invention. Amplifier 700 includes first and second input stages720 and 725, an output stage 710, first and second feedback stages 730and 735, and a feedback network 750. The amplifier 700 alsoadvantageously includes a first auto-zero circuit 740 coupled to theinput stage 720, the output stage 710, and the feedback stage 730. Theamplifier 700 further includes a second auto-zero circuit 745 coupled tothe input stage 725, the output stage 710, and the feedback stage 735.

In one embodiment, the auto-zero circuits 740 and 745 serve to switchthe amplifier 700 between first and second configurations correspondingto first and second phases of operation. For example, the firstconfiguration may correspond to an auto-zero configuration of theauto-zero circuit 740 and an amplification configuration of theauto-zero circuit 745. Conversely, a second configuration may correspondto an auto-zero configuration of the auto-zero circuit 745 and anamplification configuration of the auto-zero circuit 740.

During the first phase, the first input stage 720 and the first feedbackstage 730 are auto-zeroed while the second input stage 725 and thesecond feedback stage 735 perform the amplification functions ofamplifier 700. Conversely, during the second phase, the second inputstage 725 and the second feedback stage 735 are auto-zeroed while thefirst input stage 720 and the first feedback stage 730 perform theamplification functions of amplifier 700.

Thus, during the first phase, the auto-zero circuit 740 is operable tonull offset currents generated by the first input stage 720 and thefirst feedback stage 730. In one embodiment, the auto-zero circuit 740nulls the offset currents by shorting inputs of the first input stage720 and the first feedback stage 730 to respective common mode voltages.Subsequently, the auto-zero circuit 740 may then measure correspondingoffset currents generated by the first input stage 720 and the firstfeedback stage 730 and generate a nulling current based thereon. Thenulling current serves to compensate for the offset currents generatedby the first input stage 720 and the first feedback stage 730.

Concurrently, V_(in) is applied to the second input stage 725, V_(fb) isapplied to the second feedback stage 735, and the second input stage 725and second feedback stage 735 are coupled with the output stage 710 viathe second auto-zero circuit 745. The amplifier 700 then operatessimilar to a traditional current feedback instrumentation amplifier,except that a nulling current injected by the second auto-zero circuit745 (which is calibrated in the second phase, discussed below) ensuresthat the input-referred offset voltages of the second input stage 725and the second feedback stage 735 do not contribute to the outputvoltage.

At the end of the first phase, the first auto-zero circuit 740 changesfrom an auto-zero configuration to an amplification configuration, andthe second auto-zero circuit 745 changes from an amplificationconfiguration to an auto-zero configuration. Thereafter, the auto-zerocircuit 740 continues nulling the offset current at the outputs of thefirst input stage 720 and the first feedback stage 730.

Subsequently, in the second phase, V_(in) is applied to the first inputstage 720, V_(fb) is applied to the first feedback stage 730, and thefirst input stage 720 and the first feedback stage 730 are coupled withthe output stage 710 via the first auto-zero circuit 740. The amplifier700 then operates similar to a traditional current feedbackinstrumentation amplifier, except that the nulling current injected bythe first auto-zero circuit 740 ensures that the input-referred offsetvoltages of the first input stage 720 and the first feedback stage 730do not contribute to the output voltage.

During the second phase, while the first input stage 720 and the firstfeedback stage 730 are performing amplification functions, the secondauto-zero circuit 745 is operable to null offset currents generated bythe second input stage 725 and the second feedback stage 735. In oneembodiment, the second auto-zero circuit 745 nulls the offset currentsby shorting inputs of the second input stage 725 and the second feedbackstage 735 to respective common mode voltages. Subsequently, the secondauto-zero circuit 745 may then measure corresponding offset currentsgenerated by the second input stage 725 and the second feedback stage735 and generate a nulling current based thereon. The nulling currentserves to compensate for the offset currents generated by the secondinput stage 725 and the second feedback stage 735.

During operation, the auto-zeroing circuits 740 and 745 of amplifier 700periodically switch amplifier 700 between the first configuration andthe second configuration, ensuring that the input stages 720 and 725 andfeedback stages 730 and 735 are periodically recalibrated. Thus, this“ping-pong” operation ensures that there is continuously an offset-freestage in the signal path.

It is appreciated that amplifier 700 may be achieved in a number ofways. For example, FIG. 8 illustrates a schematic of a current feedbackinstrumentation amplifier 800, including parallel input stages, inaccordance with various embodiments of the present invention. Inamplifier 800, tranconductance amplifier 811 serves as an output stage,such as output stage 710 in amplifier 700, tranconductance amplifiers812 and 822 serve as first and second input stages, such as input stages720 and 725 of amplifier 700, tranconductance amplifiers 813 and 823serve as first and second feedback stages, such as feedback stages 730and 735 of amplifier 700, and resistors 861 and 862 together serve as afeedback network, such as feedback network 750 of amplifier 700.Although embodiments may be described herein with reference to singletranconductance amplifiers, it should be appreciated that embodimentsare not limited as such. For example, a cascade of multiple stages withappropriate frequency compensation may be used in place of the singletranconductance amplifier 811 in order to obtain a higher gain.Capacitors 851 and 852 serve as frequency compensators for thetranconductance amplifier 811, thus forming a Miller-compensated outputstage. Resistors 861 and 862, together with the reference voltageV_(ref), generate the feedback voltage V_(fb) based on the outputvoltage V_(out). V_(fb) is fed back as an input to the feedbacktranconductance amplifiers 813 and 823.

Switches 831-836 and 841-846, tranconductance amplifiers 814 and 815,and capacitors 853 and 854 function together as a first auto-zerocircuit, such as auto-zero circuit 740 of amplifier 700. Similarly,switches 871-876 and 881-886, tranconductance amplifiers 824 and 825,and capacitors 858 and 859 function together as a second auto-zerocircuit, such as auto-zero circuit 745 of amplifier 700. It should beappreciated that switches 831-836, 841-846, 871-876, and 881-886 may beany of a number of devices capable of performing a switching function.In one embodiment, the switches 831-836, 841-846, 871-876, and 881-886serve to switch the amplifier 800 between first and secondconfigurations corresponding to first and second phases of operation.For example, the first configuration may correspond to switches 831-836and 871-876 being closed and switches 841-846 and 881-886 being open.Conversely, a second configuration may correspond to switches 841-846and 881-886 being closed and switches 831-836 and 871-876 being open.

During the first phase, the first input stage and the first feedbackstage are auto-zeroed while the second input stage and the secondfeedback stage perform the amplification functions of amplifier 800.Conversely, during the second phase, the second input stage and thesecond feedback stage are auto-zeroed while the first input stage andthe first feedback stage perform the amplification functions ofamplifier 800.

Thus, during the first phase, the inputs of the tranconductanceamplifiers 812 and 813 are shorted to the input common mode voltageV_(cmin) and the feedback common mode voltage V_(cmfb), respectively.Any input offsets of amplifiers 812 and 813 cause an offset current thatflows into the integrator formed by tranconductance amplifier 814 andcapacitors 853 and 854. The output of this integrator then drives thetranconductance amplifier 815 to generate a nulling current, whicheffectively nulls the offset current.

Concurrently, V_(in) and V_(fb) are applied to amplifiers 822 and 823,respectively, and the summed output current of amplifiers 822, 823, and825 is coupled with the output stage (i.e., amplifier 811, etc.). Theamplifier 800 then operates similar to a traditional current feedbackinstrumentation amplifier, except that a nulling current injected byamplifier 825 (which is calibrated in the second phase, discussed below)ensures that the input-referred offset voltages of amplifiers 822-823 donot contribute to the output voltage.

At the end of the first phase, switches 831-836 and 871-876 open. As aresult, the voltage at the output of the integrator around amplifier 814is held so that amplifier 815 continues nulling the offset current atthe outputs of amplifiers 812 and 813.

Subsequently, in the second phase, switches 841-846 and 881-886 areclosed. V_(in) and V_(fb) are applied to amplifiers 812 and 813,respectively, and the summed current of amplifiers 812, 813, and 815 iscoupled with the output stage (i.e., amplifier 811, etc.). The amplifier800 then operates similar to a traditional current feedbackinstrumentation amplifier, except that the nulling current injected byamplifier 815 insures that the input-referred offset voltages ofamplifiers 812-813 do not contribute to the output voltage.

During the second phase, while the first input stage and the firstfeedback stage are performing amplification functions, the second inputstage and the second feedback stage are auto-zeroed. In other words, theinputs of the tranconductance amplifiers 822 and 823 are shorted to theinput common mode voltage V_(cmin) and the feedback common mode voltageV_(cmfb), respectively. Any input offsets of amplifiers 822 and 823cause an offset current that flows into the integrator formed bytranconductance amplifier 824 and capacitors 858 and 859. The output ofthis integrator then drives the tranconductance amplifier 825 togenerate a nulling current, which effectively nulls the offset current.

During operation, the auto-zeroing circuits of amplifier 800periodically switch amplifier 800 between the first configuration andthe second configuration, ensuring that the input stages and feedbackstages are periodically recalibrated. Thus, this “ping-pong” operationensures that there is continuously an offset-free stage in the signalpath.

It is appreciated that switching between parallel input and feedbackstages may cause corresponding transients to appear in the outputsignal. Therefore, in one embodiment, a high-frequency feedforward pathmay be used in combination with the ping-pong architecture of FIG. 8.FIG. 9 illustrates a block diagram of a current feedback instrumentationamplifier 900, including parallel input stages and a high-frequencyfeedforward path, in accordance with various embodiments of the presentinvention. Amplifier 900 includes first, second, and third input stages920 925, and 970, an output stage 910, first, second, and third feedbackstages 930, 935, and 980, and a feedback network 950. The amplifier 900also advantageously includes a first auto-zero circuit 940 coupled tothe input stage 920, the output stage 910, and the feedback stage 930.The amplifier 900 further includes a second auto-zero circuit 945coupled to the input stage 925, the output stage 910, and the feedbackstage 935.

In one embodiment, the auto-zero circuits 940 and 945 serve to switchthe amplifier 900 between first and second configurations correspondingto first and second phases of operation. For example, the firstconfiguration may correspond to an auto-zero configuration of theauto-zero circuit 940 and an amplification configuration of theauto-zero circuit 945. Conversely, a second configuration may correspondto an auto-zero configuration of the auto-zero circuit 945 and anamplification configuration of the auto-zero circuit 940.

During the first phase, the first input stage 920 and the first feedbackstage 930 are auto-zeroed while the second input stage 925 and thesecond feedback stage 935 perform the amplification functions ofamplifier 900. Conversely, during the second phase, the second inputstage 925 and the second feedback stage 935 are auto-zeroed while thefirst input stage 920 and the first feedback stage 930 perform theamplification functions of amplifier 900.

Thus, during the first phase, the auto-zero circuit 940 is operable tonull offset currents generated by the first input stage 920 and thefirst feedback stage 930. In one embodiment, the auto-zero circuit 940nulls the offset currents by shorting inputs of the first input stage920 and the first feedback stage 930 to respective common mode voltages.Subsequently, the auto-zero circuit 940 may then measure correspondingoffset currents generated by the first input stage 920 and the firstfeedback stage 930 and generate a nulling current based thereon. Thenulling current serves to compensate for the offset currents generatedby the first input stage 920 and the first feedback stage 930.

Concurrently, V_(in) is applied to the second input stage 925, andV_(fb) is applied to the second feedback stage 935, and the second inputstage 925 and second feedback stage 935 are coupled with the outputstage 910 via the second auto-zero circuit 945. The amplifier 900 thenoperates similar to a traditional current feedback instrumentationamplifier, except that a nulling current injected by the secondauto-zero circuit 945 (which is calibrated in the second phase,discussed below) ensures that the input-referred offset voltages of thesecond input stage 925 and the second feedback stage 935 do notcontribute to the output voltage.

At the end of the first phase, the first auto-zero circuit 940 changesfrom an auto-zero configuration to an amplification configuration, andthe second auto-zero circuit 945 changes from an amplificationconfiguration to an auto-zero configuration. Thereafter, the auto-zerocircuit 940 continues nulling the offset current at the outputs of thefirst input stage 920 and the first feedback stage 930.

Subsequently, in the second phase, V_(in) is applied to the first inputstage 920, V_(fb) is applied to the first feedback stage 930, and thefirst input stage 920 and the first feedback stage 930 are coupled withthe output stage 910 via the first auto-zero circuit 940. The amplifier900 then operates similar to a traditional current feedbackinstrumentation amplifier, except that the nulling current injected bythe first auto-zero circuit 940 ensures that the input-referred offsetvoltages of the first input stage 920 and the first feedback stage 930do not contribute to the output voltage.

During the second phase, while the first input stage 920 and the firstfeedback stage 930 are performing amplification functions, the secondauto-zero circuit 945 is operable to null offset currents generated bythe second input stage 925 and the second feedback stage 935. In oneembodiment, the second auto-zero circuit 945 nulls the offset currentsby shorting inputs of the second input stage 925 and the second feedbackstage 935 to respective common mode voltages. Subsequently, the secondauto-zero circuit 945 may then measure corresponding offset currentsgenerated by the second input stage 925 and the second feedback stage935 and generate a nulling current based thereon. The nulling currentserves to compensate for the offset currents generated by the secondinput stage 925 and the second feedback stage 935.

During operation, the auto-zeroing circuits 940 and 945 of amplifier 900periodically switch amplifier 900 between the first configuration andthe second configuration, ensuring that the input stages 920 and 925 andfeedback stages 930 and 935 are periodically recalibrated. Thus, this“ping-pong” operation ensures that there is continuously an offset-freestage in the signal path.

For low frequencies (e.g., below the clock frequency), the ping-pongauto-zeroed paths comprising input stages 920 and 925, feedback stages930 and 935, and auto-zero circuits 940 and 945 are dominant, and theamplifier 900 then operates similar to a traditional current feedbackinstrumentation amplifier, except that the nulling currents injected bythe auto-zero circuits 940 and 945 ensure that the input-referredoffsets of input stages 920 and 925 and feedback stages 930 and 935 donot contribute to the output voltage.

At high frequencies, the feedforward path comprising input stage 970 andfeedback stage 980 is dominant. Above a threshold frequency, thefeedforward path ensures that a feedback signal V_(fb) can track theinput signal V_(in). As a result, even if mixing occurs due to thegating at the inputs of input stages 920 and 925 and feedback stages 930and 935, the resulting mixing products cancel.

FIG. 10 illustrates a schematic of a current feedback instrumentationamplifier 1000, including parallel input stages and a high-frequencyfeedforward path, in accordance with various embodiments of the presentinvention. In amplifier 1000, tranconductance amplifiers 1011 and 1016together serve as an output stage, such as output stage 910 in amplifier900, tranconductance amplifiers 1012, 1022, and 1017 serve as first,second, and third input stages, such as input stages 920, 925, and 970of amplifier 900, tranconductance amplifiers 1013, 1023, and 1018 serveas first, second, and third feedback stages, such as feedback stages930, 935, and 980 of amplifier 900, and resistors 1061 and 1062 togetherserve as feedback network, such as feedback network 950 of amplifier900. Additionally, tranconductance amplifier 1016, along with capacitors1056-1057, serves as a Miller-compensated intermediate stage toamplifier 1000. Capacitors 1051 and 1052 serve as frequency compensatorsfor the tranconductance amplifier 1011, thus forming anested-Miller-compensated output stage. Resistors 1061 and 1062,together with the reference voltage V_(ref), generate the feedbackvoltage V_(fb) based on the output voltage V_(out). V_(fb) is fed backas an input to the feedback tranconductance amplifiers 1013, 1023, and1018.

Switches 1031-1036 and 1041-1046, tranconductance amplifiers 1014 and1015, and capacitors 1053 and 1054 function together as a firstauto-zero circuit, such as auto-zero circuit 940 of amplifier 900.Similarly, switches 1071-1076 and 1081-1086, tranconductance amplifiers1024 and 1025, and capacitors 1058 and 1059 function together as asecond auto-zero circuit, such as auto-zero circuit 945. It should beappreciated that switches 1031-1036, 1041-1046, 1071-1076, and 1081-1086may be any of a number of devices capable of performing a switchingfunction. In one embodiment, the switches 1031-1036, 1041-1046,1071-1076, and 1081-1086 serve to switch the amplifier 1000 betweenfirst and second configurations corresponding to first and second phasesof operation. For example, the first configuration may correspond toswitches 1031-1036 and 1071-1076 being closed and switches 1041-1046 and1081-1086 being open. Conversely, a second configuration may correspondto switches 1041-1046 and 1081-1086 being closed and switches 1031-1036and 1071-1076 being open.

During the first phase, the first input stage and the first feedbackstage are auto-zeroed while the second input stage and the secondfeedback stage perform the amplification functions of amplifier 1000.Conversely, during the second phase, the second input stage and thesecond feedback stage are auto-zeroed while the first input stage andthe first feedback stage perform the amplification functions ofamplifier 1000.

Thus, during the first phase, the inputs of the tranconductanceamplifiers 1012 and 1013 are shorted to the input common mode voltageV_(cmin) and the feedback common mode voltage V_(cmfb), respectively.Any input offsets of amplifiers 1012 and 1013 cause an offset currentthat flows into the integrator formed by tranconductance amplifier 1014and capacitors 1053 and 1054. The output of this integrator then drivesthe tranconductance amplifier 1015 to generate a nulling current, whicheffectively nulls the offset current.

Concurrently, V_(in) and V_(fb) are applied to amplifiers 1022 and 1023,respectively, and the summed output current of amplifiers 1022, 1023,and 1025 is coupled with the intermediate stage (i.e., amplifier 1016).The amplifier 1000 then operates similar to a traditional currentfeedback instrumentation amplifier, except that a nulling currentinjected by amplifier 1025 (which is calibrated in the second phase,discussed below) ensures that the input-referred offset voltages ofamplifiers 1022-1023 do not contribute to the output voltage.

At the end of the first phase, switches 1031-1036 and 1071-1076 open. Asa result, the voltage at the output of the integrator around amplifier1014 is held so that amplifier 1015 continues nulling the offset currentat the outputs of amplifiers 1012 and 1013.

Subsequently, in the second phase, switches 1041-1046 and 1081-1086 areclosed. V_(in) and V_(fb) are applied to amplifiers 1012 and 1013,respectively, and the summed current of amplifiers 1012, 1013, and 1015is coupled with the intermediate stage (i.e., amplifier 1016). Theamplifier 1000 then operates similar to a traditional current feedbackinstrumentation amplifier, except that the nulling current injected byamplifier 1015 insures that the input-referred offset voltages ofamplifiers 1012-1013 do not contribute to the output voltage.

During the second phase, while the first input stage and the firstfeedback stage are performing amplification functions, the second inputstage and the second feedback stage are auto-zeroed. In other words, theinputs of the tranconductance amplifiers 1022 and 1023 are shorted tothe input common mode voltage V_(cmin) and the feedback common modevoltage V_(cmfb), respectively. Any input offsets of amplifiers 1022 and1023 cause a corresponding offset current that flows into the integratorformed by tranconductance amplifier 1024 and capacitors 1058 and 1059.The output of this integrator then drives the tranconductance amplifier1025 to generate a nulling current, which effectively nulls the offsetcurrent.

During operation, the auto-zeroing circuits of amplifier 1000periodically switch amplifier 1000 between the first configuration andthe second configuration, ensuring that the input stages and feedbackstages are periodically recalibrated. Thus, this “ping-pong” operationensures that there is continuously an offset-free stage in the signalpath.

At high frequencies, the feedforward path comprising amplifiers1017-1018 is dominant. Together with the output amplifier 1011, it formsa regular Miller-compensated two-stage amplifier with approximately 20dB/dec roll-off. In one embodiment, the frequency at which thefeedforward path starts to dominate is:ω_(pz) =g ₁₀₁₈ /C ₁₀₅₁,  (2)(assuming C₁₀₅₁=C₁₀₅₂ and g₁₀₁₇=g₁₀₁₈). In a preferred embodiment, thisfrequency is chosen to be below the clock frequency. Above ω_(pz), thefeedforward path ensures that the feedback signal V_(fb) can track theinput signal V_(in). As a result, switching transients associated withswitching between first and second configurations are suppressed. Thelower ω_(pz), the higher the relative gain of the feedforward path atthe clock frequency and its harmonics, and therefore the better theattenuation of such switching transients.

In some cases, a residual offset may appear in amplifiers 400, 500, 600,700, 800, 900, and 1000. This residual offset may be produced by anumber of factors. For example, with reference to FIG. 4, due to thefinite output impedance of amplifiers 412, 413, and 415, the offset ofthe output amplifier 411 may result in an offset current that is notcompensated for by the auto-zeroing loop formed by amplifiers 414-415,and therefore causes an input-referred offset voltage. Second, thefinite gain in the auto-zeroing loop may result in a residualinput-referred offset voltage. Thirdly, due to charge injection at theend of the auto-zeroing phase, the voltage stored on integratorcapacitors 453-454 may change slightly, resulting in a small error inthe nulling current injected by amplifier 415 during the amplificationphase. It should be appreciated that similar effects may occur inamplifiers 500, 600, 700, 800, 900, and 1000.

In one embodiment, both residual offset due to the offset of the outputstage and the residual offset due to finite gain in the auto-zeroingcircuitry can be reduced by adding a current buffer stage 1110, as shownin FIG. 11. Although amplifier 1100 as illustrated in FIG. 11 does notinclude a feedforward path or ping-pong circuitry, it should beappreciated that the addition of a current buffer stage in a similarmanner may be achieved for amplifiers 500, 600, 700, 800, 900, and 1000.

The current buffer stage 1110 increases the impedance at the input ofamplifier 411. Therefore, the gain in the auto-zero loop is increasedand the voltage offset of amplifier 411 results in a smaller offsetcurrent. In the embodiment depicted in FIG. 11, offset introduced by thecurrent buffer stage 1110 itself is removed by the auto-zeroing process.The current buffer may be implemented as a simple cascode stage. Gainboosting may be applied to further reduce the residual offset. Inanother embodiment, an actual gain stage may be applied instead of, orin combination with, the current buffer stage 1110. It should beappreciated that in such a case, an extra dominant pole will beintroduced that will require additional frequency compensation.

In one embodiment, the residual offset due to charge injection may bekept small by using fully-differential circuitry. Charge injection wouldthen be reduced to charge-injection mismatch. In one embodiment, theoffset may be further reduced by using small switches and largeintegrator capacitors. In yet another embodiment, the transconductance(i.e., g₄₁₅, g₈₁₅, etc.) of the nulling amplifiers 415, 615, 815, 825,1015, and 1025 may be made smaller than the transconductance of theirrespective input and feedback amplifiers, so that the voltages at theoutputs of the integrators will be larger than the offset voltages atthe inputs of the respective input and feedback amplifiers. The smallerthe transconductance of the nulling amplifiers, the smaller theinput-referred offsets due to given errors in the voltages at theoutputs of the integrators.

It should be appreciated that the tranconductance amplifiers 412, 612,812, 822, 1012, and 1022 of FIGS. 4, 6, 8, and 10 have associatedamounts of input capacitance. As such, the amplifiers 412, 612, 812,822, 1012, and 1022 may act as switched-capacitor loads to the signalsource V_(in) because they are periodically discharged duringauto-zeroing phases and need to be recharged during amplificationphases. In various embodiments, this effect may be reduced by using apre-charging technique. FIG. 12 illustrates an input stage 1200 of anamplifier (such as amplifier 400) that includes pre-charging circuitry,in accordance with various embodiments of the present invention. Itshould be appreciated that similar configurations may be used inamplifiers 300, 500, 600, 700, 800, 900, and 1000 as well. In FIG. 6,the input capacitance of amplifier 412 is depicted by capacitor 1255.Input stage 1200 includes additional switches 1271 and 1272, which allowfor a “pre-charging” configuration of the input stage 1200, in additionto the amplification configuration and the auto-zeroing configuration.During the pre-charging phase, switches 431-432 and 441-442 are openedand switches 1271-1272 are closed. As a result, the inputs of amplifier412 are coupled with a buffered version of the input signal V_(in) viabuffers 1281 and 1282, so that the current needed to charge the inputcapacitance 1255 is provided by the buffer amplifiers 1281-1282, ratherthan by the signal source. In one embodiment, the buffers are 1281-1282are unity gain buffers. Thus, in a subsequent amplification phase, thesignal source only needs to provide current to correct for any smalloffset errors of the buffer amplifiers 1281-1282, rather than the fullinput voltage V_(in). It should be appreciated that while an input stage1200 is depicted in FIG. 12, the input capacitances of othertranconductance amplifiers may be pre-charged in a similar fashion. Forexample, feedback amplifiers 413, 613, 813, 823, 1013, and 1023 may bepre-charged to V_(fb) to reduce loading from their respective feedbacknetworks.

Exemplary Operations in Accordance with Various Embodiments

The following discussion sets forth in detail the operation of presenttechnology for reducing effects of offsets in current feedbackinstrumentation amplifiers. With reference to FIGS. 13-17, flowcharts1300, 1350A, 1410A, 1600, and 1625A each illustrate example operationsused by various embodiments of the present technology for reducingeffects of offsets in current feedback instrumentation amplifiers.Flowcharts 1300, 1350A, 1410A, 1600, and 1625A include processes that,in various embodiments, are carried out by circuitry in an integratedcircuit. Although specific operations are disclosed in flowcharts 1300,1350A, 1410A, 1600, and 1625A, such operations are examples. That is,embodiments are well suited to performing various other operations orvariations of the operations recited in flowcharts 1300, 1350A, 1410A,1600, and 1625A. It is appreciated that the operations in flowcharts1300, 1350A, 1410A, 1600, and 1625A may be performed in an orderdifferent than presented, and that not all of the operations inflowcharts 1300, 1350A, 1410A, 1600, and 1625A may be performed.

FIG. 13 illustrates a flowchart 1300 of a process for reducing effectsof offsets in a current feedback instrumentation amplifier, inaccordance with various embodiments of the present invention. At block1310, an input stage of the instrumentation amplifier may optionally bepre-charged with a buffered version of the input voltage prior toactually applying the input voltage to the input stage. At block 1320, afeedback stage is similarly pre-charged with a buffered version of afeedback voltage. It is appreciated that the precharge voltages may varysomewhat from the input and feedback voltages themselves. However,pre-charging in this manner reduces loading of the input and feedbackvoltages by any input capacitances in the input stage and feedback stagerespectively.

Block 1330 involves generating an intermediate current based on theinput voltage. Block 1340 involves generating a feedback current basedon an output voltage of the instrumentation amplifier. It is appreciatedthat in a conventional instrumentation amplifier, the intermediatecurrent and feedback current would have error components due to inputoffsets of the input stage and the feedback stage. Thus, at block 1350,a nulling current is generated based on the offset components. It shouldbe appreciated that generating the nulling current may be achieved in anumber of ways. For example, FIG. 14 illustrates a flowchart 1350A of aprocess for generating a nulling current, in accordance with variousembodiments of the present invention. At block 1410, the instrumentationamplifier is switched from an amplification configuration to anauto-zero configuration. It should be appreciated that this may also beachieved a number of ways. For example, FIG. 15 illustrates a flowchart1410A of a process for switching and instrumentation amplifier from anamplification configuration to an auto-zero configuration, in accordancewith various embodiments of the present invention. At block 1510, theoutput stage of the instrumentation amplifier is decoupled from theinput stage in the feedback stage. During this period while the outputstage is separated from the other stages of the amplifier, additionalcircuitry may be employed in order to effectively hold the output of theinstrumentation amplifier. At block 1520, the input stage and thefeedback stage are coupled with an auto-zero loop. This auto-zero loopmay be substantially as described and shown above, but is not limited assuch. Block 1530 then involves a common mode input voltage to the inputstage. Similarly, block 1540 involves applying a common mode feedbackvoltage to the feedback stage.

With reference again to FIG. 14, block 1420 involves measuring theoffset components. In one embodiment, this is achieved using anintegrator, but is not limited as such. The nulling current is thengenerated based on the measured offset components (block 1430). At block1440, the instrumentation amplifier is switched from the auto-zeroconfiguration back to the amplification configuration. Thus, theinstrumentation amplifier continues to compensate for the offsets duringthe amplification configuration by continuing to inject the nullingcurrent.

With reference again to FIG. 13, block 1360 involves optionallybuffering the intermediate current, the feedback current, the nullingcurrent, or any combination thereof. In one embodiment, this may beachieved through the use of a cascode stage. At block 1370, ahigh-frequency path, which may be operated concurrently with theauto-zeroed low-frequency path, is utilized to generate the outputvoltages at frequencies above a particular frequency (e.g., the clockfrequency). In other words, at high frequencies, the high-frequency pathdominates the low-frequency auto-zeroed path, and the low-frequencyauto-zeroed path dominates the high-frequency path at frequencies belowthe threshold frequency.

Referring now to FIGS. 16A-16B, flowchart 1600 illustrates anotherprocess for reducing the effects of offsets in instrumentationamplifier, in accordance with various embodiments of the presentinvention. At block 1605, a first input stage of the instrumentationamplifier is optionally pre-charged with a buffered version of the inputvoltage. Although not illustrated in flowchart 1600, a first feedbackstage of the amplifier may similarly be pre-charged with a feedbackvoltage. Again, it is appreciated that the pre-charge voltages may varysomewhat from the input and feedback voltages themselves. However,pre-charging in this manner reduces loading of the input and feedbackvoltages by any input capacitances in the first input stage and firstfeedback stage respectively. At block 1610, a first amplification pathis provided via a first sub-circuit in a first configuration of theinstrumentation amplifier. Block 1615 then involves compensating forinput offsets of the first sub-circuit with a first nulling current. Atblock 1620 the first nulling current may optionally be buffered. Thismay be achieved, for example, with a cascode stage. It is appreciatedthat other currents may be buffered in a similar manner. At block 1625,a second nulling current of a second sub-circuit of the instrumentationamplifier is calibrated. It should be appreciated that the calibrationmay be achieved in a number of ways. For example, FIG. 17 illustrates aflowchart 1625A for a process of calibrating a nulling current, inaccordance with various embodiments of the present invention. Block 1710involves applying a common mode input voltage to the input stage (i.e.,the input stage of the second sub-circuit). Block 1720 involves applyinga common mode feedback voltage to a feedback stage (i.e., the feedbackstage of the second sub-circuit). The common mode input voltage and thecommon mode feedback voltage will cause the input stage and the feedbackstage to generate currents that correspond to any offsets of the inputstage and the feedback stage. Thus, block 1730 involves measuring theoffset components of the second sub-circuit.

With reference again to FIGS. 16A-16B, block 1630 involves switching theinstrumentation amplifier from the first configuration to a secondconfiguration. In one embodiment, the switching involves switching afirst sub-circuit of the instrumentation amplifier from an amplificationconfiguration to an auto-zero configuration and switching a secondsub-circuit of the instrumentation amplifier from an auto-zeroconfiguration to amplification configuration. At block 1635, an inputstage of the second sub-circuit is optionally pre-charged with abuffered version of the input voltage. Similarly, a feedback stage ofthe second sub-circuit may be pre-charged with a buffered version of afeedback voltage.

At block 1640, a second amplification path is provided via the secondsub-circuit while the instrumentation amplifier is in the secondconfiguration. At block 1645, input offsets of the second sub-circuitare compensated for using the second nulling current that was calibratedin block 1625. At block 1650, the second nulling current is optionallybuffered, for example, using a cascode stage. Block 1655 involvescalibrating the first nulling current in the first sub-circuit while theinstrumentation amplifier is in the second configuration. In oneembodiment, the first nulling current may be calibrated as describedabove with reference to FIG. 17, but is not limited as such.

At block 1660, the instrumentation amplifier is switched from the secondconfiguration back to the first configuration. It should be appreciatedthat this process of switching between the amplification path providedby the first sub-circuit and the second sub-circuit may be repeatednumerous times during the operation of instrumentation amplifier. Suchcontinued switching allows for periodic recalibration of the nullingcurrents, which ensures that the output of the instrumentation amplifieris free of offset errors. Moreover, this ping-pong operation alsoensures that the instrumentation amplifier continually has a path frominput to output.

At block 1665, a high-frequency path, which may operate concurrentlywith the auto-zeroed low-frequency path, may be utilized to generate theoutput voltage. This path may be used, for example, at frequencies abovea threshold frequency. In one embodiment, the high-frequency path isseparate from the first sub-circuit and second sub-circuit of theinstrumentation amplifier.

Thus, embodiments provide technology allowing for instrumentationamplifiers with very low input-referred offset, low input current, andlow level spurious switching signals at the output. Moreover, someembodiments use a ping-pong architecture, which ensures that there isconstantly an offset-free stage in the signal path, and no additionaloffset is thereby introduced due to aliasing. Additionally, spurioussignals may be further reduced by adding a high-frequency feedforwardpath.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. An instrumentation amplifier comprising: an output stage forgenerating an output voltage; a first input stage for sensing adifferential input and generating a first intermediate current basedthereon; a first feedback stage coupled with said first input stage andsaid output stage, said first feedback stage for generating a firstfeedback current based on said output voltage; and an auto-zeroingcircuit coupled with said first input stage, said first feedback stage,and said output stage, said auto-zeroing circuit for generating anulling current, wherein said nulling current compensates for errors insaid first intermediate current and said first feedback currentresulting from input offsets in said first input stage and said firstfeedback stage; and a pre-charge circuit coupled with said first inputstage, wherein said pre-charge circuit is operable to charge an input ofsaid first input stage to a pre-charge voltage, wherein said pre-chargevoltage is based on said differential input, and wherein said pre-chargecircuit is operable to charge said input of said first input stage tosaid pre-charge voltage without loading said differential input.
 2. Theinstrumentation amplifier as recited in claim 1 further comprising: acurrent buffer stage coupled with said first input stage, said firstfeedback stage, said auto-zeroing circuit, and said output stage, saidcurrent buffer stage for buffering at least one of said firstintermediate current, said first feedback current, and said nullingcurrent; and wherein said current buffer stage increases an inputimpedance observed at an input of said output stage.
 3. Theinstrumentation amplifier as recited in claim 2 wherein said first inputstage, said first feedback stage, and said auto-zeroing circuitcomprises a high frequency path coupled with said output stage; andfurther comprising a high-frequency path coupled with said output stage,wherein said high-frequency path dominates said low-frequency path atfrequencies above a particular frequency, and wherein further saidlow-frequency path dominates said high-frequency path at frequenciesbelow said particular frequency.
 4. The instrumentation amplifier asrecited in claim 3 wherein said high-frequency path comprises: a secondinput stage coupled with said output stage, said second input stage forsensing said differential input and generating a second intermediatecurrent based thereon; and a second feedback stage coupled with saidoutput stage, said second feedback stage for generating a secondfeedback current based on said output voltage.
 5. The instrumentationamplifier as recited in claim 1 wherein said auto-zeroing circuit isoperable to detect offset currents corresponding to input offsets ofsaid input stage and said feedback stage generate a nulling currentbased on said detected offset currents.
 6. The instrumentation amplifieras recited in claim 5 wherein said auto-zeroing circuit comprises: anintegrator coupled with said first input stage and said first feedbackstage, said integrator being operable to detect said offset currents;and a tranconductance amplifier coupled with said integrator, saidtranconductance amplifier for generating said nulling current.
 7. Theinstrumentation amplifier as recited in claim 1 wherein saidauto-zeroing circuit comprises: a plurality of switches coupled withsaid first input stage and said first feedback stage, wherein saidplurality of switches are operable to short inputs of said first inputstage to an input common mode voltage, and wherein further saidplurality of switches are operable to short inputs of said firstfeedback stage to a feedback common mode voltage.
 8. The instrumentationamplifier as recited in claim 7 wherein said plurality of switches areoperable to temporality disconnect said first input stage and said firstfeedback stage from said output stage while said nulling current iscalibrated.
 9. The instrumentation amplifier as recited in claim 1wherein said pre-charge circuit comprises a buffer.
 10. Aninstrumentation amplifier comprising: an output stage for generating anoutput voltage; a low-frequency path coupled with said output stage,said low-frequency path comprising: an input stage for sensing adifferential input and generating a intermediate current based thereon;a feedback stage coupled with said input stage and said output stage,said feedback stage for generating a feedback current based on saidoutput voltage; and an auto-zeroing circuit coupled with said inputstage and said feedback stage, said auto-zeroing circuit for switchingsaid instrumentation amplifier between an amplification configurationand an auto-zeroing configuration, wherein said auto-zeroing circuit isoperable to detect offset currents corresponding to input offsets ofsaid input stage and said feedback stage in said auto-zeroingconfiguration, and wherein further said auto-zeroing circuit is operableto generate a nulling current based on said detected offset currents insaid amplification configuration; a pre-charge circuit coupled with saidinput stage, wherein said pre-charge circuit is operable to charge aninput of said input stage to a pre-charge voltage, wherein saidpre-charge voltage is based on said differential input, and wherein saidpre-charge circuit is operable to charge said input of said first inputstage to said pre-charge voltage without loading said differentialinput; and a high-frequency path coupled with said output stage, whereinsaid high-frequency path dominates said low-frequency path atfrequencies above a particular frequency, and wherein further saidlow-frequency path dominates said high-frequency path at frequenciesbelow said particular frequency.
 11. The instrumentation amplifier asrecited in claim 10 wherein, in said auto-zeroing configuration, saidauto-zeroing circuit is operable to de-couple said output stage fromsaid input stage and said feedback stage, to couple said input stage andsaid feedback stage with an auto-zero loop, to couple said input stagewith a common mode input voltage, and to couple said feedback stage witha common mode feedback voltage.
 12. The instrumentation amplifier asrecited in claim 10 further comprising: a current buffer stage coupledwith said input stage, said feedback stage, said auto-zeroing circuit,and said output stage, said current buffer stage for buffering at leastone of said intermediate current, said feedback current, and saidnulling current.
 13. A method comprising: generating an intermediatecurrent based on an input voltage of a current feedback instrumentationamplifier; generating a feedback current based on an output voltage of acurrent feedback instrumentation amplifier, wherein said intermediatecurrent and said feedback current comprise offset componentscorresponding to input offsets of an input stage and a feedback stagerespectively; generating a nulling current based on said offsetcomponents, wherein said nulling current compensates for said offsetcomponents; and pre-charging said input stage with a buffered version ofsaid input voltage prior to applying said input voltage to said inputstage.
 14. The method as recited in claim 13 further comprising:applying a common mode input voltage to said input stage; applying acommon mode feedback voltage to said feedback stage; and measuring saidoffset components while said common mode input voltage and said commonmode feedback voltage are applied to said input stage and said feedbackstage.
 15. The method as recited in claim 13 wherein generating saidnulling current comprises: switching said instrumentation amplifier froman amplification configuration to an auto-zero configuration; measuringsaid offset components; and generating said nulling current based on themeasuring of said offset components.
 16. The method as recited in claim15 wherein said switching comprises: de-coupling said output stage fromsaid input stage and said feedback stage; coupling said input stage to acommon mode input voltage; and coupling said feedback stage to a commonmode feedback voltage.
 17. The method as recited in claim 13 furthercomprising: buffering said intermediate current, said feedback currentand said nulling current.
 18. The method as recited in claim 13 furthercomprising: utilizing a high-frequency path to generate said outputvoltage at frequencies above a particular frequency.